Generally, multipath is formed between a base station apparatus and a communication terminal apparatus by reflection, diffraction, and/or dispersion of radio waves under the influence of buildings and suchlike. As a result, multiplied waves interfere mutually and generate multipath fading. In the HSDPA (High Speed Downlink Packet Access) scheme and suchlike known as high speed downlink packet transmission, since the modulation M-ary number for packet channels is as high as 16 QAM, reception performance will deteriorate severely by inter-path interferences resulting from the multipath fading.
Then, it is possible to carry out adaptive equalization in the communication terminal apparatus of the receiving side using a common pilot channel as a reference signal. Generally, in many cases, spreading factor is set to be large for the common pilot channel in order to enhance SNR (Signal to Noise Ratio), and set to be small for channels for high speed packet transmission in order to improve transmission efficiency.
FIG. 1 is a block diagram showing a configuration of a conventional reception apparatus. In this figure, signal transmitted from a base station apparatus is received at RF receiver 12 through antenna 11, and predetermined reception processing (a down conversion, A/D conversion and so forth) is carried out with the received signal (reception signal) at RF receiver 12. The signal after the reception processing is outputted to PL spread section 13 and adaptive equalization section 17.
At PL spread section 13, the signal outputted from RF receiver 12 after the reception processing is despread with a spreading code which is created in the base station apparatus by spreading the common pilot signal code-multiplexed with the reception signal, and the despread signal is outputted to channel estimation section 14 and correlation matrix and correlation vector calculation section 15.
Based on the despread common pilot signal outputted from PL spread section 13, at channel estimation section 14, phase variation component and amplitude variation component that the signal transmitted from the base station apparatus acquired on propagation path are estimated as channel estimation values, and the estimated channel estimation values are outputted to the correlation matrix and correlation vector calculation section 15.
At correlation matrix and correlation vector calculation section 15, an autocorrelation matrix of the input signal and a cross-correlation vector between the input signal and a desired signal are calculated, using the despread common pilot signal outputted from PL spread section 13. When these calculations are carried out, the channel estimation values outputted from channel estimation section 14, that is, the variations over the propagation path are taken into account. The calculated correlation matrix and correlation vector are outputted to weight calculation section 16.
At weight calculation section 16, the correlation matrix and the correlation vector outputted from the correlation matrix and correlation vector calculation section 15 are used to calculate an optimal weight based on the least square error criterion of the input signal vector, that is, tap coefficients for adaptive equalization section 17. The calculated weight is outputted to adaptive equalization section 17.
At adaptive equalization section 17, adaptive equalization of the signal outputted from RF receiver 12 after the reception processing is carried out using the weight outputted from weight calculation section 16. Thereby, interference components of the subject cell signal can be equalized. The signal after the adaptive equalization is outputted to packet CH spreading section 18.
The signal outputted from adaptive equalization section 17 is despread using spreading code for packet channels at packet CH spreading section 18, and then demodulated at demodulator 19. The demodulated signal is turbo-decoded at decoder 20 to obtain packet data.
Thus, the common pilot channel has a large spreading factor and can be detected as a known signal of high-quality in the reception apparatus, and so this common pilot channel can ensure the reliability of a known signal (reference signal) which is indispensable in the operation of the adaptive equalization section and improve performance of the adaptive equalization section.
Next, the basic of the calculation of the weight used at adaptive equalization section 17 will be explained. First, assuming that the number of feed-back tap and the number of feed-forward tap are L1 and L2 respectively as the tap number at adaptive equalization section 17, the input signal vector can be defined using the despread signal of the common pilot signal r(t) as follows:r(n)=[r(n−L1), . . . ,r(n−1),r(n),r(n+1), . . . ,r(n+L2)]T  (1)
Also, if a delay profile is created in interval [M1, M2], the input signal vector can be expressed by the following matrix form:r(n)=Hu(n)+n(n)  (2)
where H is a propagation path matrix having (L1+L2+1)×(L1+L2+1+M2−M1) dimension, and u is a transmitted signal sequence having (L1+L2+1+M2−M1) dimension, and H and u can be expressed as follows:
                    H        =                  [                                                                      h                  ⁡                                      (                                          M                      2                                        )                                                                                                h                  ⁡                                      (                                                                  M                        2                                            -                      1                                        )                                                                              ⋯                                                              h                  ⁡                                      (                                          M                      1                                        )                                                                              0                                            ⋯                                            0                                                                    0                                                              h                  ⁡                                      (                                          M                      2                                        )                                                                              ⋯                                                              h                  ⁡                                      (                                                                  M                        1                                            +                      1                                        )                                                                                                h                  ⁡                                      (                                          M                      1                                        )                                                                              0                                            ⋯                                                                    ⋮                                            ⋮                                            ⋮                                            ⋮                                            ⋮                                            ⋮                                            ⋮                                                                    0                                            ⋯                                            0                                                              h                  ⁡                                      (                                          M                      2                                        )                                                                              ⋯                                                              h                  ⁡                                      (                                                                  M                        1                                            +                      1                                        )                                                                                                h                  ⁡                                      (                                          M                      1                                        )                                                                                ]                                    (        3        )            andu(n)=[u(n−L1−M2), . . . ,u(n−L1−M1),u(n−L1+1−M1), . . . ,u(n+L2−M1)]  (4)
The optimal weight C based on the least square error criterion can be evaluated for this input signal vector as follows:C(n)=R(n)−1P(n)  (5)
where R represents the autocorrelation matrix (hereinafter referred to as “correlation matrix”) of the input signal, and P represents the cross-correlation vector (hereinafter referred to as “correlation vector”) between the input signal and the desired signal, and R and P can be calculated as follows:
                                          R            ⁡                          (              n              )                                =                                    ∑                              m                =                1                            n                        ⁢                                          λ                                  n                  -                  m                                            ⁢                              r                ⁡                                  (                  n                  )                                            ⁢                                                r                  H                                ⁡                                  (                  n                  )                                                                    ⁢                                  ⁢        and                            (        6        )                                          P          ⁡                      (            n            )                          =                              ∑                          m              =              1                        n                    ⁢                                    λ                              n                -                m                                      ⁢                          r              ⁡                              (                n                )                                      ⁢                                          u                1                *                            ⁡                              (                n                )                                                                        (        7        )            
In equations (6) and (7), λ is the forgetting factor, and weighted addition is carried out considering propagation variation. In addition, in Equation (7), u1 is a symbol of the common pilot channel.
Thus, if the conventional adaptive equalization section is used, reception quality can be increased by utilizing the high spreading gain for the common pilot channel to equalize interference component of the subject cell signal.
However, reception apparatus has the following problems in the above-mentioned conventional. As mentioned above, spreading factor is large for the common pilot channel in order to enhance SNR, and is small for the packet channels in order to improve transmission efficiency. Therefore, the correlation matrix and the correlation vector used in weight calculation at an adaptive equalization section are in condition in which noise and the subject cell interference component are suppressed, because of the high spreading gain of the common pilot channel. However, in the packet channels which are actual targets of equalization, since the spreading factor is lower than that of the common pilot channel, a spreading gain comparable to that of common pilot channel cannot be obtained, and noise is enhanced through operation of the adaptive equalization section. Hereinafter, more detailed explanation is given with reference to the drawings.
FIGS. 2A through 2C are conceptual diagrams showing signal components under processing in a conventional reception apparatus, and shows the sizes of the signal components schematically. FIG. 2A shows components of signal S1 outputted from RF receiver 12. Spreading suppress the common pilot channel and packet channels in size of signal component and component of the subject cell, component of other-cell and white noise occupy more than half of the whole of the components.
FIG. 2B shows the components of signal S2 outputted from PL spread section 13. It is recognized that despreading the common pilot channel of high spreading factor enlarges the components of the common pilot channel and decreases other components.
FIG. 2C shows components of signal S3 outputted from packet CH spreading section 18. It is recognized that despreading the packet channel signal obtains the spreading gain and enlarges the component of signal, and the components of the subject cell are removed by equalization. As mentioned above, although the other-cell component and white noise should be reduced by equalization, however, those components are increased by noise enhancement.
Incidentally, under the condition that noise enhancement becomes large; the demodulation performance may be inferior to that of RAKE receiver which does not mount an adaptive equalization section in the worst case.